1. Field of the Invention
The present invention relates generally to a mobile communication system, and in particular, to an apparatus and method for transmitting and receiving data in a mobile communication system using a space-time trellis code (STTC).
2. Description of the Related Art
With the rapid development of mobile communication systems, the amount of data serviced by the mobile communication system has also increased. Recently, a 3rd generation mobile communication system for transmitting high-speed data has been developed. For the 3rd generation mobile communication system, Europe has adopted an asynchronous wideband-code division multiple access (W-CDMA) system as its radio access standard, while North America has adopted a synchronous code division multiple access-2000 (CDMA-2000) system as its radio access standard. Generally, in these mobile communication systems, a plurality of mobile stations (MSs) communicate with each other via a common base station (BS). However, during high-speed data transmission in the mobile communication system, a phase of a received signal may be distorted due to a fading phenomenon occurring on a radio channel. The fading reduces amplitude of a received signal by several dB to several tens of dB. If a phase of a received signal distorted due to the fading phenomenon is not compensated for during data demodulation, the phase distortion becomes a cause of information errors of transmission data transmitted by a transmission side, causing a reduction in the quality of a mobile communication service. Therefore, in mobile communication systems, fading must be overcome in order to transmit high-speed data without a decrease in the service quality, and several diversity techniques are used in order to cope with the fading.
Generally, a CDMA system adopts a rake receiver that performs diversity reception by using delay spread of a channel. While the rake receiver applies reception diversity for receiving a multipath signal, a rake receiver applying the diversity technique using the delay spread is disadvantageous in that it does not operate when the delay spread is less than a preset value. In addition, a time diversity technique using interleaving and coding is used in a Doppler spread channel. However, the time diversity technique is disadvantageous in that it can hardly be used in a low-speed Doppler spread channel.
Therefore, in order to cope with fading, a space diversity technique is used in a channel with low delay spread, such as an indoor channel, and a channel with low-speed Doppler spread, such as a pedestrian channel. The space diversity technique uses two or more transmission/reception antennas. In this technique, when a signal transmitted via one transmission antenna decreases in its signal power due to fading, a signal transmitted via the other transmission antenna is received. The space diversity can be classified into a reception antenna diversity technique using a reception antenna and a transmission diversity technique using a transmission antenna. However, since the reception antenna diversity technique is applied to a mobile station, it is difficult to install a plurality of antennas in the mobile station in view of the mobile station's size and its installation cost. Therefore, it is recommended that the transmission diversity technique should be used in which a plurality of transmission antennas are installed in a base station.
Particularly, in a 4th generation mobile communication system, a data rate of about 10 Mbps to 150 Mbps is expected, and an error rate requires a bit error rate (BER) of 10−3 for voice, BER of 10−6 for data, and BER of 10−9 for image. The STTC is a combination of a multi-antenna technique and a channel coding technique, and is a technique bringing a drastic improvement of a data rate and reliability in a radio MIMO (Multi Input Multi Output) channel. The STTC obtains the receiver's space-time diversity gain by extending the space-time dimension of a transmitter's transmission signal. In addition, the STTC can obtain coding gain without a supplemental bandwidth, contributing to an improvement in channel capacity.
Therefore, in the transmission diversity technique, the STTC is used. When the STTC is used, coding gain having an effect of increasing transmission power is obtained together with diversity gain which is equivalent to a reduction in channel gain occurring due to a fading channel when the multiple transmission antennas are used. A method for transmitting a signal using the STTC is disclosed in Vahid Tarokh, N. Seshadri, and A. Calderbank, “Space Time Codes For High Data Rate Wireless Communication: Performance Criterion And Code Construction,” IEEE Trans. on Info. Theory, pp. 744-765, Vol. 44, No. 2, March 1998, the contents of which are incorporated herein by reference. In this reference, it is provided that if a code rate is defined as the number of symbols transmitted for a unit transmission time, the code rate must be smaller than 1 in order to obtain diversity gain equivalent to the product of the number of transmission antennas and the number of reception antennas.
FIG. 1 is a block diagram schematically illustrating a general structure of a transmitter using STTC. Referring to FIG. 1, if P information data bits d1, d2, d3, . . . , dP are input to the transmitter, the input information data bits d1, d2, d3, . . . , dP are provided to a serial-to-parallel (S/P) converter 111. Here, the index P represents the number of information data bits to be transmitted by the transmitter for a unit transmission time, and the unit transmission time can become a symbol unit. The S/P converter 111 parallel-converts the information data bits d1, d2, d3, . . . , dP and provides its outputs to first to Pth encoders 121-1 to 121-P. That is, the S/P converter 111 provides a parallel-converted information data bit d1 to the first encoder 121-1, and in this manner, provides a parallel-converted information data bit dP to the Pth encoder 121-P. The first to Pth encoders 121-1 to 121-P each encode signals output from the S/P converter 111 in a predetermined encoding scheme, and then each provide their outputs to first to Mth modulators 131-1 to 131-M. Here, the index M represents the number of transmission antennas included in the transmitter, and the predetermined encoding scheme is an STTC encoding scheme. A detailed structure of the first to Pth encoders 121-1 to 121-P will be described later with reference to FIG. 2.
The first to Mth modulators 131-1 to 131-M each modulate signals received from the first to Pth encoders 121-1 to 121-P in a predetermined modulation scheme. The first to Mth modulators 131-1 to 131-M are similar to one another in operation except the signals applied thereto. Therefore, only the first modulator 131-1 will be described herein. The first modulator 131-1 adds up signals received from the first to Pth encoders 121-1 to 121-P, multiplies the addition result by a gain applied to a transmission antenna to which the first modulator 131-1 is connected, i.e., a first transmission antenna ANT#1, modulates the multiplication result in a predetermined modulation scheme, and provides the modulation result to the first transmission antenna ANT#1. Here, the modulation scheme includes BPSK (Binary Phase Shift Keying), QPSK (Quadrature Phase Shift Keying), QAM (Quadrature Amplitude Modulation), PAM (Pulse Amplitude Modulation), and PSK (Phase Shift Keying). It will be assumed in FIG. 1 that since the number of encoders is P, 2P-ary QAM is used as a modulation scheme. The first to Mth modulators 131-1 to 131-M provide their modulation symbols S1 to SM to first to Mth transmission antennas ANT#1 to ANT#M, respectively. The first to Mth transmission antennas ANT#1 to ANT#M transmit to the air the modulation symbols S1 to SM output from the first to Mth modulators 131-1 to 131-M.
FIG. 2 is a block diagram illustrating a detailed structure of the first to Pth encoders 121-1 to 121-P of FIG. 1. For simplicity, a description will be made of only the first encoder 121-1. The information data bit d1 output from the S/P converter 111 is applied to the first encoder 121-1, and the first encoder 121-1 provides the information data bit d1 to tapped delay lines, i.e., delays (D) 211-1, 211-2, . . . , 211 -(K−1). Here, the number of the delays, or the tapped delay lines, is smaller by 1 than a constraint length K of the first encoder 121-1. The delays 211-1, 211-2, . . . , 211-(K−1) each delay their input signals. That is, the delay 211-1 delays the information data bit d1 and provides its output to the delay 211-2, and the delay 211-2 delays an output signal of the delay 211-1. In addition, input signals provided to the delays 211-1, 211-2, . . . , 211-(K−1) are multiplied by predetermined gains, and then provided to modulo adders 221-1, . . . , 221-M, respectively. The number of the modulo adders is identical to the number of the transmission antennas. In FIG. 1, since the number of the transmission antennas is M, the number of the modulo adders is also M. Further, gains multiplied by the input signals of the delays 211-1, 211-2, . . . , 211-(K−1) are represented by gi,j,t, where i denotes an encoder index, j an antenna index and t a memory index. In FIG. 1, since the number of encoders is P and the number of antennas is M, the encoder index i increases from 1 to P, the antenna index j increases from 1 to M, and the memory index K increases from 1 to the constraint length K. The modulo adders 221-1, . . . , 221-M each modulo-add signals obtained by multiplying the input signals of the corresponding delays 211-1, 211-2, . . . , 211-(K−1) by the gains. The STTC encoding scheme is also disclosed in Vahid Tarokh, N. Seshadri, and A. Calderbank, “Space Time Codes For High Data Rate Wireless Communication: Performance Criterion And Code Construction,” IEEE Trans. on Info. Theory, pp. 744-765, Vol. 44, No. 2, March 1998.
FIG. 3 is a block diagram schematically illustrating a structure of an STTC transmitter having two encoders and three transmission antennas. Referring to FIG. 3, if 2 information data bits d1 and d2 are input to the transmitter, the input information data bits d1 and d2 are applied to an S/P converter 311. The S/P converter 311 parallel-converts the information data bits d1 and d2, and outputs the information data bit d1 to a first encoder 321-1 and the information data bit d2 to a second encoder 321-2. If it is assumed that the first encoder 321-1 has a constraint length K of 4 (constraint length K=4), an internal structure, illustrated in FIG. 2, of the first encoder 321-1 is comprised of 3 delays (1+2D+D3) and 3 modulo adders, wherein the number of delays and modulo adders is equal to a value smaller by 1 than the constraint length K=4. Therefore, in the first encoder 321-1, the undelayed information data bit d1 applied to a first delay, a bit determined by multiplying a bit delayed once by the first delay by 2, and a bit delayed three times by a third delay are provided to a first modulo adder connected to a first modulator 331-1 of a first transmission antenna ANT#1. In this manner, outputs of the 3 modulo adders of the first encoder 321-1 are provided to the first modulator 331-1, a second modulator 331-2 and a third modulator 331-3, respectively. Similarly, the second encoder 321-2 encodes the information data bit d2 output from the S/P converter 311 in the same encoding scheme as that used by the first encoder 321-1, and then, provides its outputs to he first modulator 331-1, the second modulator 331-2 and the third modulator 331-3.
The first modulator 331-1 modulates the signals output from the first encoder 321-1 and the second encoder 321-2 in a predetermined modulation scheme, and then provides its output to a first transmission antenna ANT#1. It is assumed herein that a modulation scheme applied to the transmitter is QPSK. Therefore, if an output signal of the first encoder 321-1 is b1 and an output signal of the second encoder 321-2 is b2, the first modulator 331-1 modulates the output signals in the QPSK modulation scheme, and outputs b1+b2*j, where j=√{square root over (−1)}. Like the first modulator 331-1, the second modulator 331-2 and the third modulator 331-3 modulate output signals of the first encoder 321-1 and the second encoder 321-2 in the QPSK modulation scheme, and then provide their outputs to a second transmission antenna ANT#2 and a third transmission antenna ANT#3, respectively. The first to third transmission antennas ANT#1 to ANT#3 transmit to the air the modulation symbols S1 to S3 output from the first to third modulators 331-1 to 331-3, respectively.
FIG. 4 is a block diagram schematically illustrating a receiver structure corresponding to the transmitter structure of FIG. 1. Referring to FIG. 4, a signal transmitted to the air by a transmitter is received through reception antennas of the receiver. It is assumed in FIG. 4 that there are provided N reception antennas. The N reception antennas each process signals received from the air. Signals received through first to Nth reception antennas ANT#1 to ANT#N are provided to a channel estimator 411 and a metric calculator 423. The channel estimator 411 performs channel estimation on signals output from the first to Nth reception antennas ANT#1 to ANT#N, and then provides the channel estimation result to a hypothesis part 412.
A possible sequence generator 415 generates all kinds of sequences, which were possibly simultaneously encoded for information data bits in the transmitter, and provides the generated sequences to first to Pth encoders 417-1 to 417-P. Since the transmitter transmits information data by the P information bits, the possible sequence generator 415 generates possible sequences {tilde over (d)}1 . . . {tilde over (d)}P comprised of P bits. The P bits of the generated possible sequences are applied to the first to Pth encoders 417-1 to 417-P, and the first to Pth encoders 417-1 to 417-P encode their input bits in the STTC encoding scheme described in conjunction with FIG. 2, and then provide the encoded bits to first to Mth modulators 419-1 to 419-M. The first to Mth modulators 419-1 to 419-M each modulate the encoded bits output from the first to Pth encoders 417-1 to 417-P in a predetermined modulation scheme, and provide their outputs to the hypothesis part 412. The modulation scheme applied in the first to Mth modulators 419-1 to 419-M is set to any one of the BPSK, QPSK, QAM, PAM and PSK modulation schemes. Since a modulation scheme applied in the first to Mth modulators 131-1 to 131-M of FIG. 1 is 2P-ary QAM, the first to Mth modulators 419-1 to 419-M also modulate their input signals in the 2P-ary QAM modulation scheme.
The hypothesis part 412 receives signals output from the first to Mth modulators 419-1 to 419-M and the channel estimation result output from the channel estimator 411, generates a hypothetic channel output at a time when a sequence consisting of the signals output from the first to Mth modulators 419-1 to 419-M has passed the same channel as the channel estimation result did, and provides the generated hypothetic channel output to the metric calculator 423. The metric calculator 423 receives the hypothetic channel output provided from the hypothesis part 412 and the signals received through the first to Nth reception antennas ANT#1 to ANT#N, and calculates a distance between the hypothetic channel output and the signals received through the first to Nth reception antennas ANT#1 to ANT#N. The metric calculator 423 uses Euclidean distance when calculating the distance.
In this manner, the metric calculator 423 calculates Euclidean distance for all possible sequences the transmitter can transmit, and then provides the calculated Euclidean distance to a minimum distance selector 425. The minimum distance selector 425 selects a Euclidean distance having the minimum distance from Euclidean distances output from the metric calculator 423, determines information bits corresponding to the selected Euclidean distance as information bits transmitted by the transmitter, and provides the determined information bits to a parallel-to-serial (P/S) converter 427. Although there are several possible algorithms used when the minimum distance selector 425 determines information bits corresponding to the Euclidean distance having the minimum distance, it is assumed herein that a Viterbi algorithm is used. A process of extracting information bits having the minimum distance by using the Viterbi algorithm is also disclosed in Vahid Tarokh, N. Seshadri, and A. Calderbank, “Space Time Codes For High Data Rate Wireless Communication: Performance Criterion And Code Construction,” IEEE Trans. on Info. Theory, pp. 744-765, Vol. 44, No. 2, March 1998, so a detailed description thereof will not be provided for simplicity. Since the minimum distance selector 425 determines information bits corresponding to the Euclidean distance having the minimum distance for all sequences generated from the possible sequence generator 415, it finally outputs P information bits of {circumflex over (d)}1, {circumflex over (d)}1, . . . , {circumflex over (d)}P. The P/S converter 427 then serial-converts the P information bits output from the minimum distance selector 425, and outputs a reception information data sequence of {circumflex over (d)}1, {circumflex over (d)}1, . . . , {circumflex over (d)}P.
As described above, when the transmitter transmits a signal with a plurality of transmission antennas, the STTC can achieve coding gain having an effect of amplifying power of a received transmission signal, together with diversity gain, in order to prevent a reduction in channel gain occurring due to a fading channel. In the Tarokh reference, it is provided that if a code rate is defined as the number of symbols transmitted for a unit time in a communication system using STTC, the code rate must be smaller than 1 in order to obtain diversity gain corresponding to the product of the number of transmission antennas and the number of reception antennas. That is, it is provided that if it is assumed that the number of information data bits in a symbol transmitted to the air through one transmission antenna at a particular transmission time is N, even though a transmitter uses a plurality of transmission antennas, the number of information data bits that can be transmitted to the air through the plural transmission antennas at a particular transmission time must be smaller than or equal to N. The reason for providing that the number of information data bits that can be transmitted to the air through a plurality of transmission antennas should be smaller than or equal to N is to maintain diversity gain through the plural transmission antennas. Therefore, a communication system using the STTC has a difficulty in increasing its spectrum efficiency.
In addition, the communication system using the STTC has difficulty adjusting the code rate, because the code rate can be adjusted only by increasing a constellation size of modulation signals, or modulation symbols, transmitted through transmission antennas. Here, increasing a constellation size of the modulation symbols is equivalent to increasing the number of information data bits existing in each of the modulation symbols. Since it is difficult to adjust the code rate, it is impossible to adjust a code rate to a superior performance like 2.5 bits/channel use.
Finally, the communication system using the STTC has a limitation in retransmission when an error occurs in a receiver side. That is, a recently proposed wireless communication system, for example, a high-speed downlink packet access (HSDPA) communication system, provides an automatic repeat request (hereinafter referred to as “ARQ”) scheme in which when a receiver fails to normally receive a signal transmitted from a transmitter, retransmission is performed. As the ARQ scheme, an incremental redundancy (IR) scheme is typically used in which a part of the transmission-failed signal, not all of the transmission-failed signal, is retransmitted. However, the communication system using the STTC cannot use the IR scheme as the ARQ scheme because a separate puncturing scheme for a transmission signal has not been developed.